Optimal Factoring of FIR Filters

ABSTRACT

A method and system for the design and implementation of an optimally factored filter is presented. Pairs of angle values are organized in pairing candidates and a threshold is defined to indicate an upper bound on the number of pairing candidates. A first pairing candidate is exchanged above the threshold with a second pairing candidate below the threshold and a matrix is generated based on the pairing candidates below the threshold. A lowest predicted total quantization cost between all pairing candidates represented within the matrix is determined and the pairing candidates that result in the lowest predicted total quantization cost are used to determine the coefficients of the filter.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application claims the benefit of Provisional PatentApplication No. 61/941,966, filed Feb. 19, 2014, the disclosure of whichis incorporated herein by reference in its entirety.

FIELD

The invention relates generally to digital filters.

BACKGROUND

Finite impulse response (FIR) filters are commonly used digital filters.An FIR filter has an impulse response that settles to zero in a finitenumber of sample periods. FIR filters are inherently stable because FIRfilters require no feedback and have their poles at the origin (withinthe unit circle of the complex z plane). However, all digital filters,including FIR filters, are sensitive to perturbations in the filter'stap coefficients.

A digital filter constructed as a cascade of two or more sub-filters canpossess the capability of lowering the filter's sensitivity to thesefilter coefficient perturbations. This property is described in J. W.Adams and A. N. Willson, Jr., “A new approach to FIR digital filterswith fewer multipliers and reduced sensitivity,” IEEE Trans. CircuitsSyst., vol. CAS-30, pp. 277-283, May 1983 [referred to herein as“Adams”] which is herein incorporated by reference in its entirety.

A crucial capability for building such filters concerns finding the bestFIR filter factors, then carefully scaling and sequencing them. Theefficiency of the resulting structure depends heavily upon obtainingsuch optimal factors.

SUMMARY

According to an embodiment, a filter designed to receive an input signaland generate an output signal includes a plurality of first stages,where each stage of the plurality of first stages has an order of fouror greater, and one or more second stages each having an order less thanfour. The plurality of first stages and the one or more second stagesare coupled together in cascade. A total order of the plurality of firststages is higher than a total order of the one or more second stages.

According to an embodiment, a method of determining factors of a filterincludes organizing pairings of angle values into pairing candidates anddefining a threshold to indicate an upper bound on the number of pairingcandidates. The method also includes exchanging a first pairingcandidate above the threshold with a second pairing candidate below thethreshold and generating a matrix based on the pairing candidates belowthe threshold. The method then determines a lowest predicted totalquantization cost between all pairing candidates represented within thematrix and uses the pairing candidates that result in the lowestpredicted total quantization cost to determine the coefficients of thefilter.

According to an embodiment, a method for determining a sequence of aplurality of stages of a filter includes determining a sum of squaredcoefficient values for each stage of the plurality of stages of thefilter and arranging the plurality of stages of the filter in cascade.The arrangement is performed such that a first stage position in thecascade includes a stage having a lowest sum of squared coefficientvalues among each stage of the plurality of stages, and a subsequentstage position includes another stage of the plurality of stages, suchthat a partial filter comprising the another stage and each previousstage in the cascade has a lowest sum of squared coefficient valuesamong the possible stages to choose for the another stage.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

FIG. 1 depicts a zero plot for an example order-15 narrow-band low-passfilter.

FIG. 2 depicts an example cascade form (minimum order factors).

FIG. 3 depicts an FIR filter cascade, according to an embodiment.

FIG. 4 depicts an example direct-form implementation of an order-15 FIRfilter.

FIG. 5 depicts the use of carry-save adders in the 4^(th) order factors.

FIG. 6 depicts the use of carry-save adders in an example direct-formimplementation of an order-15 FIR filter.

FIG. 7 depicts a critical path in an FIR filter cascade, according to anembodiment.

FIG. 8 depicts a graph showing the required quantization level for aresulting 4^(th) order factor.

FIG. 9 depicts a zero-map and frequency response of an example 63-tapfilter having three 4th-order groups of pass-band zeros.

FIG. 10 depicts example origin & destination vectors.

FIG. 11 depicts a flow chart illustrating an example method.

FIG. 12 depicts an example of an optimally-factored FIR filter.

FIG. 13 depicts example RMS values for signals at the outputs of eachstage for the optimally factored filter.

FIG. 14 depicts an example process of filtering a signal.

FIG. 15 depicts an example illustration of partial filters in sequence.

FIG. 16 depicts example RMS of signal at outputs of a sequence of stagesof an FIR filter, according to an embodiment.

FIG. 17 depicts example RMS outputs of 60-tap filter stages, accordingto an embodiment.

FIG. 18 depicts example RMS outputs of 60-tap filter stages, accordingto an embodiment.

FIG. 19 depicts example frequency responses for all 21 stages of the60-tap filter.

FIG. 20 depicts example factors for examining truncation noise effectsat a filter output.

FIG. 21 depicts example magnitude plots of an FIR filter.

FIG. 22 depicts the complexity of a factors-pairing problem in anexample filter.

FIG. 23 depicts quantization level for a 4^(th) order factor of anexample filter, according to an embodiment.

FIG. 24 depicts a flow chart illustrating an example factoring method.

FIG. 25 depicts an example of stage insertion to reduce overallquantization cost.

FIG. 26 depicts an example magnitude response of a modifiedoptimally-factored FIR filter, according to an embodiment.

FIG. 27 depicts an example optimally factored filter, according to anembodiment.

FIG. 28 depicts a magnitude plot of a filter, according to anembodiment.

FIG. 29 depicts another magnitude plot of a filter, according to anembodiment.

FIG. 30 depicts an example optimally factored filter with fused stages,according to an embodiment.

FIG. 31 depicts a flowchart of an example method.

FIG. 32 depicts a flowchart of an example method.

FIG. 33 depicts an exemplary computer system, according to embodimentsof the present invention.

The present invention will now be described with reference to theaccompanying drawings. In the drawings, like reference numbers generallyindicate identical, functionally similar, and/or structurally similarelements. The drawing in which an element first appears is indicated bythe leftmost digit(s) in the reference number.

DETAILED DESCRIPTION OF THE INVENTION

The following detailed description of the present invention refers tothe accompanying drawings that illustrate exemplary embodimentsconsistent with this invention. References in the specification to “oneembodiment,” “an embodiment,” “an example embodiment,” etc., indicatethat the embodiment described may include a particular feature,structure, or characteristic, but every embodiment may not necessarilyinclude the particular feature, structure, or characteristic. Moreover,such phrases are not necessarily referring to the same embodiment.

Further, when a particular feature, structure, or characteristic isdescribed in connection with an embodiment, it is submitted that it iswithin the knowledge of one skilled in the art to affect such feature,structure, or characteristic in connection with other embodimentswhether or not explicitly described.

Other embodiments are possible, and modifications may be made to theembodiments within the spirit and scope of the invention. Therefore, thedetailed description is not meant to limit the invention. Rather, thescope of the invention is defined by the appended claims.

I. OVERVIEW

A digital filter has a z-domain transfer function H(z) that is arational function of the complex variable z. For linear-phase FIRfilters, the transfer function is usually put into the form of apolynomial in the variable z⁻¹ having real coefficients. Moreover, dueto the linear-phase feature, the polynomial coefficients exhibit eithereven or odd symmetry—e.g., H(z)=a+bz⁻¹+az⁻² is a filter of even order(implying an odd-length coefficient sequence) and even symmetry, whileH(z)=a+bz⁻¹−bz⁻²−az⁻³ is an example of a filter of odd order (hence, aneven length coefficient sequence) and odd symmetry. The odd-symmetrycase provides polynomials for which H(z) has a zero at dc (i.e., at z=1)and this would prohibit H(z) from being a low-pass filter. Since ourfocus here will be on low-pass filters, we shall require an Nth-orderfilter H(z) to have one of the forms given in either Equation (1a) or(1b).

H(z)=h _(N/2) +h _(N/2−1) z ⁻¹ + . . . +h ₁ z ^(−(N/2−1)) +h ₀ z ^(−N/2)+h ₁ z ^(−(N/2+1)) + . . . +h _(N/2−1) z ^(−(N−1)) +h _(N/2) z ^(−N) ,Neven  (1a)

H(z)=h _((N+1)/2) +h _((N−1)/2) z ⁻¹ + . . . +h ₁ z ^(−(N−1)/2) +h ₁ z^(−(N+1)/2) + . . . +h _((N−1)/2) z ^(−(N−1)) +h _((N+1)/2) z ^(−N) ,Nodd  (1b)

FIG. 1 shows the 15 zeros of an example type (1b) filter. This filterhas a narrow pass-band and its complex zeros appear in conjugate pairson the unit circle. One choice of factors for such a filter uses the1st-order real, and natural 2nd-order zero-pairings as factors. A factoris a polynomial component of a filter's transfer function. Such simpleminimum-order factors yield the structure shown in FIG. 2—a basic“natural” factorization.

The k-th 2nd-order complex-conjugate-pair factor of H(z) is given belowin Equation (2).

H _(k)(z)=(e ^(jθ) ^(k) −z ⁻¹)(e ^(−jθ) ^(k) −z ⁻¹)=1−(2 cos θ_(k))z ⁻¹+z ⁻².  (2)

Notice that each such 2nd-order factor has just one non-trivialcoefficient: −2 cos(θ_(k)). Such factors as shown in Equation (2) havenon-unity dc gains, which is another issue that will be discussed later.

One may also create 4th-order factors for the FIG. 1 filter by combiningtwo 2nd-order factors, which yields two non-trivial coefficients (3) forthe 4th-order factor:

$\begin{matrix}{\begin{matrix}{{{H_{p}(z)}{H_{q}(z)}} = {\left( {1 - {\left( {2\; \cos \; \theta_{p}} \right)z^{- 1}} + z^{- 2}} \right)\left( {1 - {\left( {2\; \cos \; \theta_{q}} \right)z^{- 1}} + z^{- 2}} \right)}} \\{= {1 - {2\left( {{\cos \; \theta_{p}} + {\cos \; \theta_{q}}} \right)z^{- 1}} +}} \\{{{2\left( {1 + {2\; \cos \; \theta_{p}\cos \; \theta_{q}}} \right)z^{- 2}} - {2\left( {{\cos \; \theta_{p}} + {\cos \; \theta_{q}}} \right)z^{- 3}} + z^{- 4}}}\end{matrix}\mspace{20mu} {{{First}\mspace{14mu} {non}\text{-}{trivial}\mspace{14mu} {coefficient}} = {{- 2}\left( {{\cos \; \theta_{p}} + {\cos \; \theta_{q}}} \right)}}\mspace{20mu} {{{Second}\mspace{14mu} {non}\text{-}{trivial}\mspace{14mu} {coefficient}} = {2{\left( {1 + {2\; \cos \; \theta_{p}\cos \; \theta_{q}}} \right).}}}} & (3)\end{matrix}$

An advantage to doing this combining is that the implementation of thecoefficients as shown in Equation (3) could be cheaper than theimplementation of two of the corresponding nontrivial coefficients shownin Equation (2). Moreover, there can be many possible pairings of2nd-order factors, hence many possibilities that a resulting pair ofcoefficients (3) could be particularly desirable. Let us elaborate onthis point by using the example filter of FIG. 1. In this filter thereare seven 2nd-order factors for which as many as six can be selected forpairing into three 4th-order factors. It can be shown that for N2nd-order factors, where N is even, there are 1×3× . . . x(N−1)different pairings of these N 2^(nd)-order factors. Thus, given six suchfactors (N=6), in this example, there are 3×15=15 different pairings.These 15 pairings are based upon one specific 2nd-order factor that isleft unpaired. Therefore, there are 7×15=105 different ways that one canchoose one 2nd-order factor to remain unpaired and then pair theremaining factors. This number is small enough that it would be feasibleto exhaustively explore all possibilities to find the best result aswould be understood by one of ordinary skill in the art.

However, one may consider an alternative selection of three 2nd-orderfactors to remain unpaired, which can be done in7!/(3!×4!)=5×6×7/(2×3)=35 ways. And, by pairing the remaining four2nd-order factors to make two 4th-order factors, one obtains 3×35=105additional results. Finally, if one were to select five 2^(nd)-orderfactors to remain unpaired, which can happen in 7!/(5!×2!)=6×7/2=21different ways, and pair the two remaining 2^(nd)-order factors to makea single 4th-order factor, we would have 21 additional differentresults. By adding these result totals to the one simple option ofkeeping all seven 2nd-order factors unpaired, we have 105+105+21+1=232different results.

One can perform an exhaustive search of all 232 possible pairings inthis example and find the following optimal pairings (2nd-order pair kis identified by its angle θ_(k)):

a. pair #1&#6 (27.9° & 133.25° coeffs.: −0.39774−0.42269

b. pair #2&#5 (43.7° & 110°) coeffs.: −0.76312 1.0131

c. pair #3&#7 (64.65° & 156.6° coeffs.: 0.97875 0.4284

d. leave #4)(87° unpaired. coeffs.: −0.10386

When multiplications are performed by using hard-wired shifts andadditions this best (e.g., optimal pairing) solution requires as few aseight additions, depending on whether or not certain sub-expressionreuse is employed. This is less than alternative implementations, andthis cost includes the implementation of a post-filter compensationmultiplier 302 (with binary coefficient value 0.111 shown at the filteroutput in FIG. 3). Scaling is an important topic that shall subsequentlybe discussed. It (and the closely-related topic of sub-stage sequencing)is the reason for the several (hard-wired shift) scaling multipliers 304of value 2° and 2⁻¹ along the top of FIG. 3, according to an embodiment.These scaling multipliers may be configured to shift a number of bitsassociated with the received input signal to the right if the factor oftwo is less than 1, and to shift the number of bits associated with thereceived input single to the left if the factor of two is greaterthan 1. In an embodiment, the values of these scaling multipliers arechosen to bring a DC gain of the received input signal closer to 1. Thedownward-pointing arrows that follow denote truncation of leastsignificant bits (LSBs.) The stages have been named H₁ (a 1st-orderstage), . . . , H₅. In an embodiment, each stage of the cascaded filterincludes at least two multipliers. The order of each stage (or factor)may be determined by the number of delay elements (z⁻¹) within eachstage.

As a reference point for assessing the implementation cost savings, onemay use the conventional direct-form FIR implementation of the example16-tap filter illustrated in FIG. 4. The multiplications from thisfilter require at least 15 additions. Both implementations from FIG. 3and FIG. 4 require 15 delay elements (i.e., blocks) and 15 structuraladders. However, the 4th-order stages (e.g., H₂, H₃, and H₄) in the FIG.3 structure may be organized such that they accommodate carry-savestructural adders, as shown in FIG. 5, which can yield modest increasesin the filter's operating speed and/or modest reductions in its powerconsumption.

Hard-wired shifts and additions have been mentioned as an implementationmethod and the number of such additions may be used as a measure ofimplementation cost. Another issue when comparing implementation costsis the issue of data-path word-length. It is quite possible that, as theinput data flows through a filter structure, it may be necessary for theword-length to grow. This matter can be closely related to the amount ofround-off noise that a system introduces. In the case of the examplesystem just discussed with reference to FIG. 3, it can be shown that theoptimally factored implementation of this filter system requires nogrowth in data-path word-length. We also observe (in FIG. 6) that, whilethe direct-form FIR filter can also benefit from the use of carry-saveadders, its power consumption (also given in FIG. 6) can still exceedthat of the FIG. 3 filter. Thus, it is evident that the performance ofthe optimally-factored FIR filter (as illustrated in FIG. 3) has thepotential to surpass that of the direct-form structure, both in terms ofspeed and power consumption. This performance edge is especiallyenhanced by the fact that the conventional direct-form structure cantend to require adders of greater bit-width (than those employed by theoptimally factored FIR filter) in performing the additions at the bottomof FIG. 4 or FIG. 6. Also, the notion of pipelining these additions forhigh-speed operation can become a severe power-consumption drawback forthe conventional direct-form FIR filter. Although FIG. 3 illustrates anexample FIR filter, the methods and structures described herein may alsobe applied to an IIR filter, or any other type of digital or analogfilter. According to an embodiment, the optimally factored filterillustrated in FIG. 3 includes a plurality of filters each having anorder of four or greater, as well as one or more filters each having anorder less than four. In this example, stages H2, H3, and H4 each havean order of four, while stage H1 and stage H5 have orders of one andtwo, respectively. A total order of the plurality of first stages (12 inthis example) is higher than a total order of the one or more secondstages (3 in this example). In another embodiment, none of the stageswithin the plurality of first stages are identical to one another.

A clear advantage that the optimally factored-FIR filter possesses isthat it can very easily be pipelined so as to increase its maximumoperating speed. Consider the example multi-factor structure of FIG. 3and envision its relatively long critical path, which is emphasized inFIG. 7. It is evident that, for sufficiently high-order filters, thisdelay-free path will become the overall structure's critical path. Itslength can set the filter's maximum operating speed, and this applieswhether or not the stages employ carry-save adders in the mannerillustrated in FIG. 5. The conventional direct-form FIR filter of eitherFIG. 4 or FIG. 6 will also encounter such difficulties when high-speedoperation is desired. The traditional remedy for such speed-limitingproblems is to introduce some form of pipelining into the system.

In the case of the optimally factored FIR filter, FIG. 7 shows anexample of how, by inserting two delay blocks (z⁻¹) into the FIG. 7structure, the critical path delay is reduced to that of just two4th-order stages, according to an embodiment. In another example,registers may be used in place of the two delay blocks (z⁻¹). The delayblocks or registers may be considered to be examples of pipeliningbuffers. The number of such insertions and their locations may bedetermined so as to create a system having the minimum number ofpipelining registers that provide a desired high operating speed. Thepipelining of the conventional direct-form FIR filters tends to be muchmore difficult and hardware-costly in comparison to theiroptimally-factored FIR counterparts.

As noted above, the performance of the optimally-factored FIR filter hasthe potential to surpass the direct-form structure both in terms ofspeed and power consumption. The next section will discuss how optimalfactors may be obtained.

II. SETTING UP THE ALGORITHM

According to an embodiment, the method for determining optimal FIRfilter factors begins by obtaining a rough indication of the qualitiesof the 4^(th)-order transfer function factors that can be made from two2^(nd)-order factors. Here, the 2nd-order factors may be associated withthe complex-conjugate stop-band zeros of the sort shown in FIG. 1, but2nd-order factors obtained from reciprocally-valued real zeros may alsobe included and may be present in a low-pass FIR filter's transferfunction. For such factors we have a transfer function as shown below inequation (4):

H _(k)(z)=(a−z ⁻¹)(a ⁻¹ −z ⁻¹)=1−(a+a ⁻¹)z ⁻¹ +z ⁻²  (4)

FIG. 8 illustrates the approximate required quantization level (i.e.,the number of bits, excluding the sign bit) that would be needed torepresent the resulting 4th-order-factor coefficients when pairing two2nd-order factors having angles θ_(p) and θ_(q) represented as points onthe x and y axes, respectively. (Angle values are given in degrees, with0.5° resolution.) The determined values for θ_(p) and θ_(q) may be usedto determine the filter coefficients for each stage as shown in Equation(3). While the plot shows that there are various special angles forwhich different results apply, several general features include: (1) Theplot is invariant to an exchange of the x and y axes; (2) the upper-leftand lower-right quadrants [one angle large, one small] tend to providelow quantization sensitivity levels; (3) the upper-right and lower-leftcorners [both angles large or both small] tend to yield high sensitivityquantization levels; and (4) there is a thin “border” around the entireplot for which rather low sensitivity results tend to occur. This wouldseem to correspond to one (or both) angles being close to 0° or 180°,and similarly for many angle-pairs where one angle (or both) are closeto 60°, 90°, or 120°. Further details regarding the approximate requiredquantization levels with angles θ_(p) and θ_(q) can be found in A.Mehrnia and A. N. Willson, Jr., “Optimal factoring of FIR filters,” IEEETransactions on Signal Processing, vol. 63, no. 3, pp. 647-661, Feb. 1,2015.

One may create several examples that illustrate some of the tendenciespredicted by FIG. 8. The first example considers pairing two 2nd-orderfactors with angles of θ_(p)=48° and θ_(q)=138°; this results in a verylow sensitivity level for the non-trivial coefficients of this 4th-orderfactor. Indeed, a mere 3-bit quantization level yields a frequencyresponse with negligible error. Alternatively, when pairing two2nd-order factors having angles θ_(p)=45° and θ_(q)=63°, a much highersensitivity level results; these 4th-order-factor coefficients require8+1 bits. Finally, the importance of the FIG. 8 sensitivity assessmentsis further highlighted when considering that, according to FIG. 8, thepairing of θ_(p)=13° and θ_(q)=20°, should require more than 11 bits.

We are beginning to discuss a search algorithm whose goal is to findoptimal factors for a given low-pass FIR transfer function. Unlike someof the examples discussed in the Overview, it is usually very difficultto exhaustively search through all of the possible choices for gettinggood factors. This is because the number of choices can become quitelarge. Consider the general situation with a narrow-band low-pass filterfor which there are N/2 zero pairs that can be candidates for achievinga set of best-matched pairs (4th-order factors). It can be shown thatthe number of distinct ways that one can pair together N/2 2nd-orderfactors into N/4 4th-order factors is:

(N/2−1)(N/2−3)(N/2−5) . . . 5×3×1  (5)

For each of these possible pairings, a 1-bit, 2-bit, . . . , to as largeas perhaps 32-bit quantization of the factor coefficients may needexamination for each resulting set of N/4 4th-order factors (each withas many as two non-trivial coefficients). Neglecting that there areadditional cases to be considered, i.e., finding those 2nd-order factorsthat are better left uncombined, and assuming that, on average, tenquantization possibilities per factor must be examined (out of thevarious 1-bit to 32-bit quantization possibilities per factor), thiswould represent a total of at least:

10(N/4)(N/2−1)(N/2−3)(N/2−5) . . . 5×3×1 cases  (6)

To provide some insight into the size of this number, for N=52, thisnumber exceeds 10¹⁵, and for N=36, it exceeds 3 billion. Smaller valuesof N provide more encouraging realities: for N=24 it is 623,700 and forN=12 it is 450. This, unfortunately, indicates that an impractical levelof complexity may be at hand if one is considering an exhaustiveexamination of all pairing possibilities. While the previous example ofan order-15 transfer function may have left the impression that anexhaustive search is feasible, commonly used filter sizes can often betoo large for this.

The data illustrated in FIG. 8 may be used to predict the requiredquantization levels for the 4th-order factors that result from thepairing of any two 2nd-order H(z) factors whose roots lie on the unitcircle in the z-plane, according to an embodiment. The information fromwhich FIG. 8 was created is contained in a table (a “coarse quantizationcost matrix”) that is used by the FIR filter optimal-factoring algorithmto get a good starting point for its processing. Getting a good startthis way can help to reduce the computational complexity to a morepractical range. As mentioned, the data in this table are indexed byapproximations to angles θ_(p) and θ_(q) that have a 0.5° precision. Asimilar approach is applicable for the case of pass-band zeros that lieoff the unit circle, typically being associated with pass-band ripples,as shown in FIG. 9, which displays three groups of four such zeros, eachgroup being described by a 4th-order polynomial factor taking the form:

$\begin{matrix}{\begin{matrix}{{H_{k}(z)} = {\left( {{r_{k}^{{j\theta}_{k}}} - z^{- 1}} \right)\left( {{r_{k}^{- {j\theta}_{k}}} - z^{- 1}} \right)\left( {{r_{k}^{- 1}^{- {j\theta}_{k}}} - z^{- 1}} \right)\left( {{r_{k}^{- 1}^{{j\theta}_{k}}} - z^{- 1}} \right)}} \\{= {\left( {r_{k}^{2} - {\left( {2\; r_{k}\cos \; \theta_{k}} \right)z^{- 1}} + z^{- 2}} \right)\left( {r_{k}^{- 2} - {\left( {2\; r_{k}^{- 1}\cos \; \theta_{k}} \right)z^{- 1}} + z^{- 2}} \right)}} \\{= {1 - {2\left( {r_{k} + r_{k}^{- 1}} \right)\cos \; \theta_{k}z^{- 1}} + {\left( {r_{k}^{2} + r_{k}^{- 2} + {4\; \cos^{2}\theta_{k}}} \right)z^{- 2}} -}} \\{{{2\left( {r_{k} + r_{k}^{- 1}} \right)\cos \; \theta_{k}z^{- 3}} + z^{- 4}}}\end{matrix}\mspace{20mu} {{{First}\mspace{14mu} {non}\text{-}{trivial}\mspace{14mu} {coefficient}} = {{- 2}\left( {r_{k} + r_{k}^{- 1}} \right)\cos \; \theta_{k}}}\mspace{20mu} {{{Second}\mspace{14mu} {non}\text{-}{trivial}\mspace{14mu} {coefficient}} = {r_{k}^{2} + r_{k}^{- 2} + {4\; \cos^{2}{\theta_{k}.}}}}} & (7)\end{matrix}$

Even when such zeros exist, we shall tend to focus on pairing the2nd-order unit-circle-zero factors.

III. DYNAMIC SEARCH ALGORITHM

With regards to the 16-tap filter example of FIG. 1, Table 1 below showsthe estimated quantization cost of the various pairings of the filter'sseven 2nd-order factors, according to an embodiment. Since we would notpair one of these factors with itself, the entries on the main diagonalin Table 1 are understood to represent the cost for leaving each pairalone.

TABLE 1 Course Quantization - Number of bits needed for all pairs angle27.9° 43.7° 64.65° 87° 110° 133.25° 156.6° zero pair #1 #2 #3 #4 #5 #6#7 27.9° #1 9 8 7 6 6 4 5 43.7° #2 8 8 6 7 3 2 4  64.65° #3 7 6 6 6 5 44 87°   #4 6 7 6 4 5 6 5 110°   #5 6 3 5 5 4 8 8 133.25° #6 4 2 4 6 8 410 156.6°  #7 5 4 4 5 8 10 6

Also, since pairing, for example, zero-pair #1 with zero-pair #2 is thesame as pairing zero-pair #2 with zero-pair #1, the cost entries inTable 1 would always constitute a symmetric matrix and would thereforecontain redundant information.

According to an embodiment, Table 1 is reorganized by sorting each rowin ascending order of cost. This will ruin the matrix symmetry, so eachrow entry is accompanied with the column-name from which it originallycame. This yields the result shown below in Table 2, for which eachentry has a cost, above the cell's diagonal line, and the originalcolumn number below (where #k denotes column k).

TABLE 2 Sorted Entries on Rows From Table 1

The cells that represent redundant information have been indicated bycrosshatching (these are data that originated in cells below the maindiagonal of Table 1). The first element of the fifth row of Table 2shows that pairing the 2nd-order factor #5 with the 2nd-order factor #2results in a 4th-order factor with two non-trivial coefficients, as inEquation (3), that require an estimated three bits, that is 3+1,including a sign bit. Notice that this cell of Table 2 is one of theredundant cells—its information is already provided by the second entryin row 2.

As noted Table 2, zero-pair #3 may be paired with either #6 or #7 toresult in the lowest approximate quantization cost. Also, 2nd-orderfactor #4 is observed to require the least quantization cost when it isnot paired. These tentative conclusions are based on the initial coarsequantization cost matrix and hence the algorithm will need to examinethese attractive pairing choices more precisely in its next phases.

As shown in Table 3 below, a “depth” parameter can also be defined; itindicates an upper bound on the number of potential pairing candidatesthat will be considered for all rows. For instance, a depth value offour indicates that at most the first four candidates in each row of thesorted coarse quantization cost matrix will be considered during therest of the optimization process. (In addition, all self-pairingpossibilities will always be considered, as indicated in FIG. 10 by thepresence of a path from node #k to node #k, for all k.) While such adepth parameter may seem unnecessary for smaller filters with limitednumbers of zero pairs, it represents an efficient, virtually essentialsimplifying procedure when dealing with very large filters.

TABLE 3 Slight Rearrangement of Row elements from Table 2

According to an embodiment, one more alteration to Table 2 addresses thefact that, for example, a crosshatched cell (#2\6) on row 3 of the tablecan be exchanged with an uncrosshatched cell (#4\6) while not increasingthe cost (both have 6 as the cost) and that will give one additionalpairing option while keeping the depth-4 setting. Similarly, cells #1\6and #6\6 can be exchanged on row 4. These exchanges are incorporatedinto Table 3. Even beyond this alteration, the algorithm will, in fact,always include any non-redundant cell that has a cost not greater thanthe highest cost appearing to the left of the depth barrier.

With the depth-4 pairings identified in Table 3 expressed as origin anddestination nodes above and below the arrows in FIG. 10, a binaryconnection matrix (shown below in Table 4) may be created that containsthe feasible flow information (allowed pairings) allowing for thedetermination of a binary flow (solution) vector x that will indicatethe optimal pairings, according to an embodiment. This information canbe put into matrix and vector form as Ax, where, for the depth-4 choiceof the present example, the vector x is a length-22 vector containingonly binary (0 and 1) entries and the 7×22 matrix A, given in equation(8), conveys the information of Table 4.

TABLE 4 Binary Connection Matrix (Feasible Pairings) depth = 4 originvector 1 1 1 1 1 2 2 2 2 2 3 3 3 3 3 4 4 4 4 5 6 7 destination vector 14 5 6 7 2 3 5 6 7 3 4 5 6 7 4 5 6 7 5 6 7 #1 1 1 1 1 1 0 0 0 0 0 0 0 0 00 0 0 0 0 0 0 0 #2 0 0 0 0 0 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 #3 0 0 00 0 0 1 0 0 0 1 1 1 1 1 0 0 0 0 0 0 0 #4 0 1 0 0 0 0 0 0 0 0 0 1 0 0 0 11 1 1 0 0 0 #5 0 0 1 0 0 0 0 1 0 0 0 0 1 0 0 0 1 0 0 1 0 0 #6 0 0 0 1 00 0 0 1 0 0 0 0 1 0 0 0 1 0 0 1 0 #7 0 0 0 0 1 0 0 0 0 1 0 0 0 0 1 0 0 01 0 0 1

$\begin{matrix}{A = \begin{bmatrix}1 & 1 & 1 & 1 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 1 & 1 & 1 & 1 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 1 & 1 & 1 & 1 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 1 & 1 & 1 & 1 & 0 & 0 & 0 \\0 & 0 & 1 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 1 & 0 & 0 & 1 & 0 & 0 \\0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 1 & 0 & 0 & 1 & 0 \\0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 1 & 0 & 0 & 1\end{bmatrix}} & (8)\end{matrix}$

The vector x may be determined such that Ax=b holds, where vector b haslength 7 with all “1” elements. Since each column in matrix A representsa specific zero pairing, the length-22 vector x will have at-most seven“1” entries and the rest will be “0.” That is, x will select up to sevencolumns of A and, since each column in this selection will represent apairing of two 2nd-order factors (or it will represent the choice that afactor will remain unpaired), for each row of A there will be exactlyone selected column that has a non-zero value on that row. It is thealgorithm's task to determine the vector x that specifies the optimalchoice of these columns, according to an embodiment.

Binary Integer Programming provides one example tool to use in solvingproblems specified in the following form: Find the binary vector x thatminimizes a linear (scalar product) function ƒ^(T)x subject to thelinear constraints Ax≦b, where x is binary (i.e., x is a vector of 0 and1 values only). In conventional linear programming, the Ax≦binequalities can be made to include equality constraints (Aeq x=beq) andthus such constraints can even become the only constraints of interestfor certain applications. In this case, to employ binary integerprogramming, the scalar product ƒ^(T)x may be used to include thevarious costs of the pairings of the 2^(nd)-order factors—e.g., theupper numbers in the cells of Table 3.

Thus, for our example (and with depth 4), the required quantizationcosts for all possible pairings are one more than (to include the signbit) each number in this sequence: [9 6 6 4 5 8 6 3 2 4 6 6 5 4 4 4 5 65 4 4 6]. To correctly represent the total quantization cost, anyunpaired zero-pair results in a 2^(nd)-order factor with one non-trivialcoefficient, as shown in equation (2), while any actual pairing yields a4th-order factor with two non-trivial coefficients, as shown in equation(3), hence requiring twice the quantization cost. A quadratic costweighting may be used, although linear, cubic, etc., could also beemployed. These modifications lead to the cost vector: ƒ^(T)=[100 98 9850 72 81 98 32 18 50 49 98 72 50 50 25 72 98 72 25 25 49]. One or moreentries of the cost vector may be related to one or more quantizationcosts among the pairing candidates.

For this example problem, the binary integer programming tool must findthe binary vector x satisfying Ax=b, such that ƒ^(T)x is minimized,where A is given by equation (8), ƒ is given by the cost vector shownabove, and where b is a length-7 vector in which each entry is “1.” Toavoid creating a bias that could possibly favor one potential pairingchoice, an x_(o) is used that leaves all zero pairs alone (i.e., ifx=x_(o), then no 2nd-order factors would be paired). Thus, for thisexample, x_(o) is given as: x_(o)=[1 0 0 0 0 1 0 0 0 0 1 0 0 0 0 1 0 0 01 1 1]^(T). Given that Table 1 provides a coarse quantization costmatrix (also illustrated in FIG. 8) to initiate this design, the staticbinary integer programming problem is modified into a dynamic version inwhich the cost vector is continually fine-tuned to represent the actualcost.

FIG. 11 illustrates a flowchart 1100 for using binary integerprogramming to help determine a lowest quantization cost forimplementing a filter, according to an embodiment. The Binary Integerprogramming (BIP) model is set up at block 1102 as described, forexample, in Section II. The BIP model is then executed at block 1104. Atblock 1106, suggested pairings are generated as discussed with referenceto Tables 2 and 3. At block 1108, a lowest quantization level of theresulting factors is determined by solving, for example, for the binaryvector x satisfying Ax=b, such that ƒ^(T)x is minimized.

At block 1110, a determination is made whether the current factorsresult in an actual quantization cost that is substantially the same asthe predicted quantization cost determined in block 1108. If the actualquantization cost is substantially the same as the predictedquantization cost, then the method ends. However, if the actualquantization cost is not substantially the same as the predictedquantization cost, method 1100 proceeds to block 1112 where the costvector x is updated using the actual quantization cost values. The loopthen repeats starting at block 1104, until the actual quantization costis substantially the same as the predicted quantization cost.

According to one embodiment, as a final step, to check for havingperhaps reached a local optimum, one can launch the above-describeddynamic pairing algorithm a second time using a different depthparameter. When comparing the resulting solutions, the ultimate solutionmay be chosen as the one with the lower quantization cost. It is alsopossible to run the algorithm with even more depth values.

When the algorithm is used on the order-15 (FIG. 1) example filter, thealgorithm provides the same result as is presented in FIG. 3, accordingto an embodiment. It is also possible, for example, to include a pair ofzeros having reciprocal real values within the realm of second-orderfactors that are subject to pairing into 4th-order factors. While thesame algorithm can also be employed for filters having complex zerosthat are off the unit circle—which, due to a desire for linear-phasefilters, would naturally occur as groups of four zeros that obeyconjugate-reciprocal requirements (as shown in FIG. 9) for zeros thatproduce pass-band ripples, it is often suitable to simply representthese zeros by their natural 4th-order factors that have the four zerosas roots. It may be useful to combine two such factors to make an8th-order factor, or to combine such a 4th-order factor with another4th-order factor that was created from four unit-circle zeros with thealgorithm. Normally there is a relatively limited handful of such zerosfor a sharp narrow-band low-pass filter and they may be manually pairedwith another 2nd-order or 4th-order factor.

IV. SCALING AND STAGE SEQUENCING

To ensure a DC gain of unity for each stage in the sequence, acompensation multiplier is employed at the output of each stage,according to an embodiment. The value of this multiplier can becalculated by setting z=1 in equations (2), (3), (4), or (7), dependingon the type of stage, (and getting its reciprocal). This is efficientlyrealized by approximating such stage multipliers by the nearestpower-of-two, which may be built computation free using a hard-wiredleft-shift or right-shift of the data, according to an embodiment. Tocreate the overall scale factor, a residual scaling by the value β islumped into one post-filter compensation multiplier. FIG. 12 shows thisresulting structure.

A practical optimally factored FIR filter may require carefulfactor-sequencing to efficiently manage the data-path word-length. Theobjective here includes finding a proper sequence for which the samesuitably-small data-path word length N will suffice for all stages. Toillustrate, consider a 60-tap filter example having a best 21 factorsfound using the algorithm described in Section III. Table 5 below liststhese factors and the stages are initially sequenced (Stage #1 is first,. . . , Stage #21 is last).

TABLE 5 60 Tap Filter - Quantized stages of best identified pairedfactored sequence vs. CSD filter Best identified paired factors (directfactoring of the 60-tap filter) abs quantized (binary) Stage 60-tap CSDfilter Quantized 18 stages non-trivial coeffs Mult. Quantizedcoefficients  1. 1 + z⁻¹ none    0.5 h(0) = 2⁻⁸ − 2⁻¹³  2. 1 + z⁻² none   0.5 h(1) = 2⁻⁸ − 2⁻¹¹  3. 1 − 2.0374755859375z⁻¹ + z⁻²10.0000100110011 −32 h(2) = 2⁻⁸ − 2⁻¹¹ + 2⁻¹³  4. 1 − 0.96484375z⁻¹ −0.46875z⁻² − 0.96484375z⁻³ + z⁻⁴ [0.11110111 0.01111000]  −2 h(3) = 2⁻⁹− 2⁻¹¹ + 2⁻¹²  5. 1 − 1.8046875z⁻¹ + z⁻² 1.1100111    4 h(4) = 2⁻¹⁰  6.1 + 0.0546875z⁻¹ − 1.21875z⁻² + 0.0546875z⁻³ + z⁻⁴ [0.0000111 1.00111]   1 h(5) = −2⁻⁹ − 2⁻¹³  7. 1 − 0.21875z⁻¹ − 0.5z⁻² − 0.21875z⁻³ + z⁻⁴[0.0011 none]    1 h(6) = −2⁻⁷ + 2⁻¹¹ + 2⁻¹³  8. 1 − 0.8125z⁻¹ + 0.75z⁻²− 0.8125z⁻³ + z⁻⁴ [0.1101 0.1100]    1 h(7) = −2⁻⁶ + 2⁻⁹ − 2⁻¹¹  9. 1 +0.125z⁻¹ − 0.375z⁻² + 0.125z⁻³ + z⁻⁴ [none 0.011]    0.5 h(8) = −2⁻⁶ −2⁻⁷ + 2⁻¹¹ 10. 1 − 0.015625z⁻¹ + 0.21875z⁻² − 0.015625z⁻³ + z⁻⁴[0.000001 0.001110]    0.5 h(9) = −2⁻⁵ − 2⁻⁹ + 2⁻¹¹ 11. 1 − 1.1875z⁻¹ +z⁻² 1.0011    1 h(10) = −2⁻⁴ + 2⁻⁶ + 2⁻⁸ 12. 1 + 0.71875z⁻¹ + 0.25z⁻² +0.71875z⁻³ + z⁻⁴ [0.10111 none]    0.5 h(11) = −2⁻⁴ + 2⁻⁷ + 2⁻⁹ 13. 1 +1.125z⁻¹ + 0.375z⁻² + 1.125z⁻³ + z⁻⁴ [1.001 0.011]    0.25 h(12) =−2⁻⁴ + 2⁻¹⁰ + 2⁻¹¹ 14. 1 − 0.25z⁻¹ + 1.75z⁻² − 0.25z⁻³ + z⁻⁴ [none 1.11]   0.5 h(13) = −2⁻⁴ − 2⁻⁹ − 2⁻¹¹ 15. 1 − 0.4375z⁻¹ + z⁻² 0.0111    0.5h(14) = −2⁻⁴ − 2⁻⁹ − 2⁻¹³ 16. 1 − 0.21875z⁻¹ + z⁻² 0.00111    0.5 h(15)= −2⁻⁴ + 2⁻⁷ − 2⁻⁹ 17. 1 + 0.1875z⁻¹ + z⁻² 0.0011    0.5 h(16) = −2⁻⁵ −2⁻⁷ − 2⁻⁹ 18. 1 + 0.625z⁻¹ + z⁻² 0.101    0.5 h(17) = −2⁻⁶ − 2⁻¹³ 19.1 + z⁻¹ + z⁻² none    0.25 h(18) = 2⁻⁶ + 2⁻⁸ − 2⁻¹² 20. 1 + 1.90625z⁻¹ +z⁻² 1.11101    0.25 h(19) = 2⁻⁴ + 2⁻⁹ − 2⁻¹² 21. 1 + 1.9921875z⁻¹ + z⁻²1.1111111    0.25 h(20) = 2⁻³ − 2⁻⁷ + 2⁻¹¹ Post-filt mult =1.180419921875 Post-filt mult = h(21) = 2⁻³ + 2⁻⁴ − 2⁻⁷ 1.0010111000110h(22) = 2⁻² − 2⁻⁸ − 2⁻¹⁰ h(23) = 2⁻² + 2⁻⁴ + 2⁻⁹ h(24) = 2⁻¹ − 2⁻³ + 2⁻⁷h(25) = 2⁻¹ − 2⁻⁴ + 2⁻⁷ h(26) = 2⁻¹ + 2⁻⁹ − 2⁻¹² h(27) = 2⁻¹ + 2⁻⁵ + 2⁻⁶− 2⁻¹² h(28) = 2⁻¹ + 2⁻⁴ + 2⁻⁶ + 2⁻¹³ h(29) = 2⁻¹ + 2⁻⁴ + 2⁻⁵ + 2⁻¹¹Post-filt mult = 0.0010001111101

A “Four-Test Procedure,” measuring the RMS value at each stage's outputmay be run on this data, according to an embodiment. The input RMS maybe 2^(N-PAPR) ^(—) ^(margin) in all four test cases. N is theword-length and setting PAPR_margin=2 accommodates an extremely highpeak-to-average power ratio (PAPR) of 12 dB. (AlternativelyPAPR_margin=1.5 supports 9-dB PAPR which can satisfy many applications.)FIG. 13 illustrates the resulting normalized RMS values for twodifferent tests. The Four-Test Procedure involves the following fourtests:

Test 1) The input signal is white Gaussian noise (uniform power acrossall frequencies). The filter is expected to attenuate by 60 dB theportion of the signal within the stop-band.

Test 2) The input signal is colored Gaussian noise with uniform powerwithin the stop-band. It is a sum of 100 random phase sinusoidsuniformly distributed across the stop-band. A 60 dB attenuation of theentire signal is expected.

Test 3) The input signal is one sinusoid at the pass-band edge.

Test 4) The input signal is one sinusoid at the stop-band edge.

The RMS results in FIG. 13 indicate high levels of overshoot (as high as65 dB) for performing Tests 1 and 2 of the Four-Test Procedure, whichwould require very large bit-width for the data-path (as much as 10 or11 extra bits) to avoid signal clipping. This illustrates why properlysequencing the stages, to manage and minimize the data-path word-length,is important.

The simple M! trial-and-error approach for stage sequencing isimpractical since each Four-Test Procedure would have to be reconfiguredM! times. A more practical stage sequencing approach is needed. First,however, suppose a stop-band attenuation of −20 log₁₀(δ_(s)) dB isdesired. FIG. 14 shows a situation where a post-A/D signal must have itsout-of-band noise removed, according to an embodiment. Thesignal-to-quantization-noise ratio is given below in Equation (9).

$\begin{matrix}{{SQNR}_{{filter}\mspace{20mu} {input}} = {\frac{\left. {{Power}({signal})} \right)}{{Power}\left( {{quant}\mspace{14mu} {noise}} \right)} = {\frac{\left( 2^{N - {PAPR\_ margin}} \right)^{2}}{\left( {0.5 - (00.5)} \right)^{2}/12} = {\left. {12\left( 2^{N - {PAPR\_ margin}} \right)^{2}}\Rightarrow{{SQNR}({dB})} \right. = {{10\; {\log_{10}\left( {12\left( 2^{N - {PAPR\_ margin}} \right)^{2}} \right)}} - {6\left( {N - {PAPR\_ margin}} \right)} + 10.8}}}}} & (9)\end{matrix}$

This is also the level of quantization noise that is injected bytruncation at the output of each stage in FIG. 12. Given the targetstop-band attenuation of −20 log₁₀(δ_(s)) dB it is important to reducethe overall effect of all stage quantization noise to a substantiallynegligible level compared to the stop-band attenuation. This indicatesthat we need a sufficiently large “margin” and that Equation (10) belowsatisfied:

SQNR(dB)≧−20 log₁₀(δ_(s))+margin(dB)  (10)

With M optimal-factor stages and N+1 bits (including the sign bit) inthe signal path, as in FIG. 12, and considering that the stagequantization noise sources are independent and identically distributed(iid), the required word-length is given by Equation (11):

$\begin{matrix}{\mspace{79mu} \left. {{{SQNR}({dB})} \geq {{{- 20}\; {\log_{10}\left( \delta_{s} \right)}} + {{margin}({dB})} + {10\; {\log_{10}(M)}}}}\Rightarrow{N \geq {\frac{{- 10.8} - {20\; {\log_{10}\left( \delta_{s} \right)}} + {{margin}({dB})} + {10\; {\log_{10}(M)}}}{6} + {PAPR\_ margin}}} \right.} & (11)\end{matrix}$

Inequality (11) provides a useful relationship between the word lengthN, the value of the margin, and the PAPR_margin. It may facilitate aniterative process to get the desired small (yet large enough) value ofN. Thus Equation (11) indicates that for (−60 dB) and M=21 (e.g., Table5), and using PAPR_margin=1.5, the signal path word-length (N),excluding sign bit, should be at least 13 bits to support a margin of atleast 6 dB (to limit stop-band degradation) for a majority applications.Also, if the input x(t) to a linear system with an impulse response h(t)is stationary white noise with power P, then the output y(t) power canbe described by Equation (12):

E[y ²(t)]=rms _(y) ² =P∫ _(−∞) ^(+∞) |h(t)|² dt=P∫ _(−∞) ^(+∞) |H(ƒ)|²df.  (12)

Equation (12) shows that, to minimize the increases in data-path RMS anddynamic range at the stage outputs, it is important to determine thespecific sequence of stages that would minimize the sum of squaredcoefficient values for all partial filters—where partial filters aredefined as in FIG. 15, according to an embodiment. This observationenables a simple sequential approach in sequencing the M optimal factorswithout the need to, each time, reconfigure the complete sequence fortest. According to an embodiment, for the first position in thesequence, out of the M possible choices, choose the factor with thesmallest sum-of-squared coefficient values (recall, all stages haveapproximately 0-dB DC gain). Next, for the second position, out of theM-1 remaining possible stages, choose the one that yields the smallestsum-of-squared coefficient values for the partial filter #2 asillustrated in FIG. 15. Then, for the third position, out of the M-2remaining stages, choose the one that yields the smallest sum-of-squaredcoefficient values for the resulting partial filter #3 as illustrated inFIG. 15. This process is continued until the last stage in the sequenceis reached, according to an embodiment. This entails a more practicalcomplexity level of M+(M−1)+ . . . +2=M(M+1)/2−1, which is 230 for theexample where M=21. The sequence of RMS values for Tests 1 and 2 arereported in FIG. 16 for the resulting sequence of optimal factors.

While, as shown in FIG. 16 (for the example of the 60-tap filter with 21stages), this proposed initial sequencing method is very effective inmitigating and fully resolving the RMS inflation problem shown in FIG.13; unfortunately, as one remaining draw-back, it also forces one of thefactors with the largest sum-of-squared coefficient values to be thelast stage (Factor #3 for this 60-tap filter example). This is alsoillustrated in FIG. 16, where the very last stage (Factor #3)excessively amplifies its input noise and the residual stop-band signalto an unacceptable level at its output (which is the filter's output)and hence, prevents the overall filter from achieving the target 60-dBstop-band attenuation.

To address this final issue, according to an embodiment, a simplemodification of the preliminary sequencing result is adopted. The “worstfactors” are identified in terms of the previously introduced metric.Table 6 shows factors with the largest sum-of-squared coefficient valueswhen all factors are normalized to have a DC gain of 0 dB, according toan embodiment. A specific threshold for the largest sum-of-squaredcoefficient values may be used (e.g., threshold=4) to define the numberof worst factors. Other worst factor (WF) settings could be used aswell. Table 6 gives the features of the worst eight factors for thepresent example. The WF selected factors are then distributed atisolated fixed positions that spread across the sequence of optimalfactors, according to an embodiment.

TABLE 6 Worst factors of the 60-tap filter example optimal factorsLargest Worst Factors (top to bottom) among Table 5 stages Sum ofabsolute (But when normalized to DC gain of 1, we have, for squaredvalue of example −26.68 + 54.37z⁻¹ − 26.68z⁻² for Factor #3) coeffs.coeffs. 3. 1 − 2.0374755859375z⁻¹ + z⁻² 4379.9 54.37 5. 1 −1.8046875z⁻¹ + z⁻²  137.8  9.24 4. 1 − 0.96484375z⁻¹ − 0.46875z⁻² − 25.7  2.51 0.96484375z⁻³ + z⁻⁴ 11. 1 − 1.1875z⁻¹ + z⁻²   5.2  1.46 6.1 + 0.0546875z⁻¹ − 1.21875z⁻² +   4.4  1.37 0.0546875z⁻³ + z⁻⁴ 8. 1 −0.8125z⁻¹ + 0.75z⁻² − 0.8125z⁻³ + z⁻⁴   3.1  0.94 7. 1 − 0.21875z⁻¹ +0.5z⁻² − 0.21875z⁻³ + z⁻⁴   2.1  0.89 15. 1 − 0.4375z⁻¹ + z⁻²   0.9 0.64

Next the above-described process is repeated, generating the sequence ofpartial filters illustrated in FIG. 15, once for each possiblepermutation of the WF selected worst factors, according to anembodiment. This final sequencing algorithm has a total complexity of:[M(M+1)/2−1](WF!)=16,200 for M=21−5. The resulting “best identifiedsequence” for the 60-tap filter example is: Stage Order=[13 10 8 6 14 94 12 17 7 11 20 15 3 21 16 2 5 19 18 1]. The results of the Four-TestProcedure are shown in FIG. 17 for Test 1 and 2 of the Four-TestProcedure, and in FIG. 18 for Tests 3 and 4 of the Four-Test Procedure,according to an embodiment. The five worst factors are circled to showtheir enforced positions in the sequence. The best permutation is [6 411 3 5] for these enforced positions in this example. Both FIG. 17 andFIG. 18 show that the “best identified sequence” is able to fullyattenuate (by at least 60 dB) the stop-band portion of the input signal(including a sinusoid positioned at the edge of the stop-band) and it isable to pass the pass-band signal (including a sinusoid positioned atthe edge of the pass-band) with negligible (less than 0.1 dB)attenuation. The frequency responses of all 21 stages, ordered accordingto the identified sequence are shown in FIG. 19.

Another example of checking the behavior of the designated stagesequence may be employed, to exhibit the effects of quantization noisecaused by truncations at stage-outputs. Taking a smaller example, suchas the order-15 filter illustrated in FIG. 3, the noise appearing at theoutput of the filter that is due to the truncation operation at theoutput of each of its five stages may be accounted for. Working from theoutput backwards, the first truncation encountered appears directly atthe output, moreover, the next truncation, appearing directly at theoutput of Stage H5, is immediately connected to the filter output afterbeing scaled by the output scaling factor β=7/8, according to anembodiment. For the other truncation-noise sources, those at the inputsof stages H2, H3, H4, and H5, the transfer functions may be defined asG1, . . . , G4 (via their impulse responses) as follows: h₁=[1 1]×0.5;h₂=[1−0.3984375−0.421875−0.3984375 1]×1; h₃=[1−0.75 1−0.75 1]×0.5; h₄=[11 0.4375 1 1]×0.5; h₅=[1−0.09375 1]×0.5. Note that performing atruncation is one method for signal datapath wordlength (bitwidth)management. Another method includes rounding instead of truncation.

The G1, . . . , G4 partial-filters as illustrated in FIG. 20 are:g₁=h₅×β; g₂=conv(h₄, g₁); g₃=conv(h₃, g₂); g₄=conv(h₂, g₃), according toan embodiment. Example magnitude plots for these transfer functions areshown in FIG. 21. These G1, . . . , G4 partial-filters provide nosignificant amplification of their input quantization noise and hencedemonstrate that the FIG. 3 stages would not require data-pathword-length growth. In the instance when insufficient stop-bandattenuation or pass-band “transparency” occurs one can also resort to asmall increase in the data-path word-length (above the value N) just forthe most troublesome stage(s).

V. EXTENSION OF OPTIMAL FACTORING ALGORITHM

The general problem of optimally pairing factors associated withunit-circle zeros, as discussed above, addresses the realization of alower complexity optimally factored cascade structure. A closed-formexpression approximating the total number of different choices of zeropairings for the case of M factors may be found as:

$\begin{matrix}\left\{ \begin{matrix}{{\approx {\frac{M!}{\sqrt{2^{M}}{\left( \frac{M}{2} \right)!}}\cosh \; \sqrt{M}}},} & {{if}\mspace{14mu} M\mspace{14mu} {is}\mspace{14mu} {Even}} \\{{\approx {\sqrt{M}\frac{\left( {M - 1} \right)!}{\sqrt{2^{M - 1}}{\left( \frac{M - 1}{2} \right)!}}\sinh \; \sqrt{M - 1}}},} & {{if}\mspace{14mu} M\mspace{14mu} {is}\mspace{14mu} {{Odd}.}}\end{matrix} \right. & (13)\end{matrix}$

FIG. 22 illustrates that an impractical order of complexity can resultif an exhaustive examination of (all possible) zero-pairing combinationsis attempted. For example, a filter with 45 natural factors (2nd- or4th-order, corresponding to on- and off-unit-circle zeros) entails atotal pairing complexity of approximately 10³⁰. Also, for each pairingpossibility, the resulting filter is formed and then the smallestcoefficient quantization level for each resulting factor must beidentified while satisfying the given filter specification, in order tobenchmark the outcome versus other pairing possibilities. Hence,identifying the optimal factoring is a problem with an overall order ofcomplexity that is approximately an order of magnitude higher than thatwhich is indicated in equation (13) and FIG. 22 (due to the additionaltask of minimum-quantization level identification).

According to an embodiment, a recursive binary integer programmingalgorithm described here extends the algorithm described previously inSection III, in that the optimization process now includes (in additionto the 2nd-order factors employed before) all 4th-order factors formedfrom off-unit-circle zeros (in general, any remaining factors) usuallythose associated with a filter's pass-band ripples. The binaryconnection matrix A contains all feasible flow information required tofind a binary flow vector x that will ultimately identify the optimalfactors of the filter. The binary vector x has length K, where K is thetotal number of possible non-redundant flows (allowed pairings) for eachfactor that is considered in the algorithm. The size of binary matrix Ais M×K where M is the number of factors that are obtained by decomposingthe given filter. The vector x may be determined such that the costfunction (ƒ^(T)x) is minimized and is subject to the linear constraintAx=b, where, the length-K vector b has all “1” elements and the elementsof the length-K cost vector ƒ are estimated costs associated with thevarious quantization levels for each of the K allowed pairings (i.e.,flows). In an embodiment, an 8th-order factor has twice, and four times,the cost of a 4th-order and a 2nd-order factor, respectively, at thesame quantization level.

Having formulated the problem, the main challenges remaining are thedefining of the coarse (initial) values of the quantization costs(elements of the vector J) and then the recursive updating of thosecosts with finer values, to deal with the nonlinear relationship betweena factor's coefficient-quantization and the overall filter's magnituderesponse. According to an embodiment, the final outcome of the algorithmis the binary vector x that has M entries with value “1” (and the restwill be “0”), in a way that for each row of the M×K matrix A there isexactly one selected column that has a non-zero value on that row. Eachcolumn in this selection will represent a pairing of two factors, or itwill represent the choice that a factor will remain unpaired.

To better explain the quantization cost associated with a 4th-orderoff-unit-circle factor, a sensitivity analysis with respect to itsmagnitude and angle is insightful. As mentioned, these zeros, typicallyassociated with pass-band ripples, appear in reciprocalcomplex-conjugate groups of four (See FIG. 22), described by a 4th-orderpolynomial factor having the form:

H _(k)(z)=(r _(k) e ^(jθ) ^(k) −z ⁻¹)(r _(k) e ^(−jθ) ^(k) −z ⁻¹)(r _(k)⁻¹ e ^(−jθ) ^(k) −z ⁻¹)(r _(k) ⁻¹ e ^(jθ) ^(k) −z ⁻¹)=1+z ⁻⁴−2(r _(k) +r_(k) ⁻¹)cos θ_(k)(z ⁻¹ +z ⁻³)+(r _(k) ² +r _(k) ⁻²+4 cos²θ_(k))z⁻²  (14)

-   -   where 0<r_(k)<1, and 0<θ_(k)<180°

Eq. (14) provides a first non-trivial coefficient of −2(r_(k)+r_(k)⁻¹)cos θ_(k) and a second non-trivial coefficient of r_(k) ²+r_(k) ⁻²+4cos²θ_(k). A suitable quantization level may be identified for the twonontrivial coefficients to employ the corresponding off-unit-circle4th-order factor in the construction of an optimally factored filter.Otherwise the frequency-response deviation caused by the quantizedfactor's implementation could violate the overall targettransfer-function specification. This deviation may be expressed interms of either the Root Mean Square Error (RMSE) or the Mean AbsoluteError (MAE). Here we employ the normalized MAE, as defined in Equation(15) where Num denotes the number of frequencies at which the functionsare evaluated across the spectrum [0 2π]. For instance, MAE levels of −2and −2.5 (in log terms) indicate normalized average error levels of 1%and 0.3% respectively for the quantized versus ideal magnitude responseof the off-unit-circle factor.

$\begin{matrix}{{{{MAE}\mspace{14mu} \left( {{in}\mspace{14mu} \log \mspace{14mu} {terms}} \right)} = {{\log_{10}\left( {\frac{1}{Num}{\sum\limits_{p = 0}^{{Num} - 1}\; {\frac{{{H_{k}\left( ^{j\frac{2\pi \; p}{Num}} \right)}} - {{{\hat{H}}_{k}\left( ^{j\frac{2\pi \; p}{Num}} \right)}}}{{H_{k}\left( ^{j\frac{2\pi \; p}{Num}} \right)}}}}} \right)}.\mspace{20mu} {Where}}},{{H_{k}\left( ^{j\frac{2\pi \; p}{Num}} \right)} = {1 - {2\left( {r_{k} + r_{k}^{- 1}} \right)\cos \; \theta_{k}^{{- j}\frac{2\pi \; p}{Num}}} + {\left( {r_{k}^{2} + r_{k}^{- 2} + {4\; \cos^{2}\theta_{k}}} \right)^{{- j}\frac{4\pi \; p}{Num}}} - {2\left( {r_{k} + r_{k}^{- 1}} \right)\cos \; \theta_{k}^{{- j}\frac{6\pi \; p}{Num}}} + ^{{- j}\frac{8\pi \; p}{Num}}}}} & (15)\end{matrix}$

and

${\hat{H}}_{k}\left( ^{j\frac{2\pi \; p}{Num}} \right)$

is the quantized version of

${H_{k}\left( ^{j\frac{2\pi \; p}{Num}} \right)}.$

FIG. 23 illustrates the approximate (coarse) quantization levels(excluding sign-bit) as a function of Magnitude (specified to within0.01 resolution) and Angle (specified to within 0.25° for representingthe first and second coefficients, such that the normalized MAE fromEquation (15) is limited to 0.3% (which represents a negligible level ofnormalized error), according to an embodiment. The plot illustratesseveral important general features:

(1) The plot is symmetric with respect to the angle of 90°.

(2) The regions having magnitude less than 0.6, independent of angle,and magnitude greater than 0.6 for angles between approximately 45° and)135° tend to provide low quantization-sensitivity results.

(3) The upper-right and lower-right corners (small angle or large angleand near-unity magnitude) tend to yield high quantization-sensitivityresults.

Notice the very large quantization levels (16+1 bits or higher) requiredfor the sensitive portion of the plot. FIG. 23 illustrates that aresulting factored filter could conceivably benefit considerably fromthe optimal pairing of an off-unit-circle zero group with another factorwhen zeros lying within the sensitive region of the plot are present.

The pairing of 4^(th) order factors may be repeated to identify theelements of the coarse (initial estimate) quantization cost vector ƒassociated with the pairing of each off-unit-circle zero group withother factors obtained from decomposing the target filter. This wouldcomplete the formulation of the binary programming problem and then wecan solve for the vector x following the same procedure as discussedabove in Section III, by using bintprog recursively in Matlab toiteratively update the cost vector with finer quantization cost values,according to an embodiment. Alternatively, we can start with unpairedfactors (obtained from decomposing the filter) to try to determine thelowest quantization level for each factor, subject to the resultingfilter meeting the target spec. It is then possible to use the outcomesas the initial coarse quantization costs (elements of vector J) for thepairing of the 4th-order factors with other factors, by scaling thecosts for the three cases of 4th-order paired with 4th-order, or leftalone, 4th-order paired with a 2nd-order, and a 2nd-order alone. Therecursive binary programming process may then be employed as describedearlier with the additional step of repeating the process a few timesfor a few cases of the cost scaling. The complete flowchart for thedescribed extended optimal factoring methodology is shown in FIG. 24.

FIG. 24 illustrates a method 2400 of determining optimal factors for afilter, which includes a series of steps 2402 associated with running apairing algorithm, according to an embodiment. Steps 2402 are similar tothose shown in FIG. 11 and described above in Section III. According tomethod 2400, once the best pairing has been achieved for the givenparameters, step 2404 determines if this is the final choice ofparameters to use. If not, then step 2406 defines a new set ofparameters and re-runs the optimization process defined in steps 2402.If the user or program is done changing the parameters, then step 2408determines if the cost matrix used is the final choice. If not, then anew cost matrix is defined in step 2412 and the optimization processdefined in steps 2402 is re-run again. If, at step 2408, the final costmatrix is determined to have been used, then the lowest quantizationcost pairings are used in step 2410. As noted in step 2410, the stagescan then be scaled, sequenced, etc., to further optimize thefunctionality of the filter.

VI. ADDITIONAL TECHNIQUES AND BENEFITS

As illustrated in Table 6 (and also in FIG. 19), certain unpaired2nd-order factors (e.g., Factor #3) can be difficult to deal with; henceit may be desirable to ensure that such a factor will always be pairedwith some other factor, according to an embodiment. One technique fordoing this is to set to zero (or omit) the relevant column from matrix A(from Equation 8). For example, by removing column 11 from A in equation(8), a binary integer program will be forced to pair zero-pair #3 withone of: #2, #4, #5, #6, or #7. In the same way, a specific pairing maybe excluded, to avoid creating an undesirable 4th-order factor, bysetting to zero column 12 in equation (8) to ensure that zero-pair #3will not be combined with zero-pair #4.

The optimal factoring algorithm provides the freedom to imposeadditional constraints that can rule out such results, and it will thenfind the best pairings for which such constraints are met, according toan embodiment. In addition to the equality constraint Ax=b, the binaryinteger programming tool can simultaneously accommodate a set ofinequality constraints Cx≦d, for which C=I, i.e., C is the K×K identitymatrix (K is the length of solution-vector x), and then, if the normaldesign is used (without such additional constraints) all elements of thevector d may be set to 1. That works because whatever “best solution”vector x one might determine via the Ax=b constraint, withoutintroducing the Cx≦d constraint, that same “best x” will also satisfyCx≦d because C=I, and x≦d will always hold, since x is a vector of 0 and1 elements, while all elements of d have the value 1. But when certainundesirable pairings are wanted, certain elements of d are set to 0. Forexample, using the matrix A provided above in Equation (8), the matrixhas seven rows corresponding to the seven factors of the filter. Thefive “1” elements in the first row indicate that Factor #1 is eitherleft unpaired (the first element) or it is allowed to be pairedaccording to either of the following factor pairings: (#1-#4), (#1-#5),(#1-#6) and (#1-#7). Suppose it is desired to ensure that Factor #1 andFactor #4 are never paired in the so-constrained optimal solution. Thenwe need only set the second element of the d vector to be 0, becausethen the Cx≦d constraint will ensure that the second element of thesolution vector x will be 0, hence it will not have the value 1 when xis chosen to satisfy Ax=b, according to an embodiment. Thus, thesolution vector x will not indicate (as the second column of A “offers”)that Factors #1 and #4 should be paired together.

More generally, multiple columns of A may simultaneously be eliminatedfrom consideration this way, by setting multiple (corresponding)elements of d to zero. In this regard, if we were to set only elements#2, #3, #4, and #5 of d to zero that would rule out the pairing ofFactor #1 with any other factor.

This provides a way to improve the prospects of getting a goodsequencing result, or it could even be viewed as a way to perhaps “giveup” a little in terms of minimizing hardware and/or power consumption toget a sub-optimally factored cascade system with better noisecharacteristics through achieving a “more tamed” magnitude response forcertain factors.

A variation on the above technique for blocking or ensuring certainpairings (while not involving directly the Cx≦d inequality constraints),is to simply set to zero (or omit) the relevant column from the flowmatrix A. For example, consider the above matrix A in Equation (8) for afilter with seven factors considered for potential pairing (each rowrepresents one factor). By setting column #11 of A to zero (it containsthe #3 unpaired choice), the optimal factoring algorithm will pairfactor #3 with one of: #2, #4, #5, #6, or #7 (enabled by the remaining“1” elements in row #3). In the same way, a specific pairing may beexcluded to avoid creating an undesirable 4th-order factor: Settingcolumn #12 to all zeros will ensure that factor #3 will not be combinedwith factor #4.

Section III described embodiments for finding the “best” factoring of anFIR transfer function, however, one may also be interested in findingthe second-best, third-best, . . . , etc., sets of optimal factors.According to an embodiment, in addition to the equality constraintsAx=b, that were used previously, the binary integer programmingalgorithm can simultaneously accommodate a set of inequality constraintsCx≦d, for which we now specify a C matrix and d vector. The originalalgorithm described in Section III may first be used to obtain theoptimal factors. Next, the vector x representing that solution may beset as a length-K column-vector, which is renamed as x_(old), accordingto an embodiment. Matrix C may then be defined to be the 1×K matrixwhose single row is just x_(old) ^(T). Additionally, M_(old) may bedefined as the following non-negative integer: M_(old)=one less than thenumber of “1” elements in x_(old), according to an embodiment. In thiscase, the Cx≦d constraint may be included in a second running of thebinary integer programming algorithm where the same A matrix and b valueis used, but we also include Cx≦d with the above 1×K matrix being C andthe above M_(old) as the single component of the length-1 vector d,according to an embodiment. The Cx≦d constraint imposes the requirementon the new solution vector x (for the second running of the binaryinteger programming algorithm) that it yield a different solution vectorx (different than x_(old)). While Ax=b holds, it follows that x≠x_(old)is equivalent to Cx≦d, which is the same as <x_(old),x>≦M_(old). Thisoccurs for two reasons:

(1) for no binary vector x can the scalar product <x_(old),x> be greaterthan M_(old)+1 since there are exactly M_(old)+1 nonzero elements inx_(old); and

(2) this scalar product can be as large as M_(old)+1 only if x=x_(old)

Taking this concept one step further, the binary integer programmingalgorithm may be run over and over, each time creating an x_(old1) andM_(old1), then x_(old2) and M_(old2), x_(old3) and M_(old3), etc. Newrows are then added to matrix C using x_(old2), x_(old3), . . . , andnew elements to a growing column-vector d as shown below in Equation(13), where N_(old)=M_(old):

$\begin{matrix}{C = {{\begin{bmatrix}X_{{old}\; 1}^{T} \\X_{{old}\; 2}^{T} \\X_{{old}\; 3}^{T}\end{bmatrix}\mspace{31mu} d} = \begin{bmatrix}N_{{old}\; 1} \\N_{{old}\; 2} \\N_{{old}\; 3}\end{bmatrix}}} & (13)\end{matrix}$

According to an embodiment, this procedure allows for the generation ofa collection of “very good” sets of optimal factors, among which one mayexpect at least one to be attractive in regard to how well its factorsscale and sequence—or, perhaps, to have other desirable properties.

An example attractive property of the optimally factored structure isits capability to accommodate further combining of its stages, or theinsertion of additional stages and the fusing of them with an existingstage in the structure without affecting any other stages. This mayresult in lower coefficient sensitivity, better overall frequencyresponse, and better noise and stage performance, according to anembodiment. For example, filter complexity (in terms of quantizationcost) may be reduced through the insertion of the new stage (1+z⁻¹) andthen “fusing it” with the most expensive stage (the second stage) of theoptimally-factored 16-tap example from FIG. 3. The convolution and newquantization from this example are illustrated in FIG. 25, and thebetter frequency response, in FIG. 26.

In another example, an order 59 filter, such as the one illustrated inFIG. 27 having 21 total factors, may include fused factors where one hasa costly coefficient (for example, a first factor having 15+1 bitquantization cost and a second factor having 6+1 bit quantization cost)to result in a new 4^(th) order factor having a lower hardwarecomplexity in comparison with the total for the previous factors if theyhad been left separated. If the rest of the 19 factors remain intact,the resulting magnitude response for the (tamed) modified factoredcascade, shown in FIG. 28, has a passband ripple that is limited to±0.1085 dB and a stopband attenuation that is better than the required60 dB specification. The resulting 4th-order factor has two non-trivialcoefficients that require only (8+1)-bit quantization, compared to, forexample, the (15+1) bit quantization of factor 1 and (6+1) bitquantization of factor 2.

The filter illustrated in FIG. 27 may be an optimally factored filterhaving a plurality of first stages where each stage within the pluralityhas an order of four or greater. Also, the filter includes one or moresecond stages each having an order less than four. A total order of theplurality of first stages may be higher than a total order of the one ormore second stages.

In this example, the number of shift-add operations needed to implementthe resulting 4th-order factor is less than the non-fused case by threeshift-adds. The magnitude response of the new 4th-order factor isplotted in FIG. 29 and it shows that the frequency domain behavior ofthe first factor (top dashed line) is considerably tamed (i.e., it hasless out-of-band gain) when it is fused (convolved) with the secondfactor (bottom dashed line). Given that the fusion of the aforementionedfactors results in the removal of two 2nd-order factors and theintroduction of the new 4th-order factor, the total number of cascadefactors is reduced by one. Thus, according to an embodiment, stagesequencing is performed again for the modified cascade. The sequencingalgorithm described above in Section IV may be run again for theremaining 20 factors. The revised optimally factored structure is shownin FIG. 30. The new 4th-order factor is identified with the bold arrow.The green dashed line indicates the portion of the cascade where thestage ordering has changed, compared to the original filter cascadeillustrated in FIG. 27. Both partial and fully pipelined versions of thefactored structure in FIG. 30 may be realized by inserting registers(buffers) before the stage inputs. This pipelining is yet one furtherbenefit of the factored structure.

VII. EXAMPLE METHODS OF OPERATION

FIG. 31 illustrates a method 3100, according to an embodiment. Themethod may be used to determine optimal coefficients of an FIR filter,IIR filter, digital filter, or analog filter. The method may beimplemented in software and executed on a computing device, orimplemented using hardware components.

Method 3100 begins at block 3102 where angle value pairings (e.g., θ_(p)and θ_(q)) are organized into pairing candidates, according to anembodiment. An example of this organization is shown in Table 2 asdescribed above in Section III. The angle value pairings may beorganized from a lowest quantization cost to a highest quantizationcost, per pairing. In one example, the quantization cost for a givenpairing of angle values is associated with a number of bits required torepresent the given pairing of angle values within a stage of thefilter. The number of bits may, or may not, include a sign bit.

Next, at block 3104, an upper threshold bound is defined, according toan embodiment. This may be the depth parameter discussed in Section IIIwhich determines the extent of the factors to be considered in theproceeding computations.

At block 3106, one or more pairing candidates above the threshold areexchanged with one or more pairing candidates below the threshold,according to an embodiment. An example of this re-organization is shownin Table 3 as described above in Section III. A quantization cost ofeach of pairing candidate exchanged with another pairing candidate maybe the same, according to an embodiment.

At block 3108, a matrix is generated based on the pairing candidatesbelow the chosen threshold, according to an embodiment. An examplematrix is provided as Matrix A as discussed above in Section III.

At block 3110, the pairing candidates within the matrix that result in alowest total quantization cost are determined, according to anembodiment. This may be performed, for example, using binary integerprograming and the process depicted in either FIG. 11 or FIG. 24. Thebinary integer programming algorithm may be used to determine a binaryvector x that minimizes a linear cost function.

At block 3112, the pairing candidates that result in the lowest totalquantization cost are used to determine the coefficients of an FIR, IIR,digital, or analog filter, according to an embodiment. If a given anglevalue pairing is represented by θ_(p) and θ_(q), then the first andsecond coefficients of the filter may be provided as −2·(cos θ_(p)+cosθ_(q)) and 2·(1+2·cos θ_(p)·cos θ_(q)), respectively.

In an embodiment, blocks 3104 through 3112 are repeated with a differentthreshold determined in block 3106 during each iteration.

After all coefficients have been determined, the resulting cascadedfilter may include stages with an order of four or higher, whileincluding one or more other stages having an order less than four. In anembodiment, the total order of the stages having an order of four orhigher is greater than a total order of the one or more stages having anorder less than four.

FIG. 32 illustrates a method 3200, according to an embodiment. Themethod may be used to determine an optimal sequence for multiple stagesof an FIR, IIR, digital, or analog filter. The method may be implementedin software and executed on a computing device, or implemented usinghardware components.

Method 3200 begins at block 3202 where the sum of squared coefficientvalues are determined for each filter stage, according to an embodiment.This is described, generally, in Section IV. In one example, more thanhalf of the stages each have an order of four or greater. In anotherexample, a subset of the stages each have an order of four or greater,and the total order of the subset of the stages is greater than a totalorder of a remainder of the stages.

At block 3204, the filter stages are arranged in cascade such that thesum of squared coefficient values among all the stages is minimized,according to an embodiment. This is also described in Section IV. Forexample, for the first stage position in the sequence, out of the Mpossible choices, choose the stage with the smallest sum-of-squaredcoefficient values. Next, for the second position, out of the M-1remaining possible stages, choose the one that yields the smallestsum-of-squared coefficient values for the partial filter #2 asillustrated in FIG. 15. Then, for the third position, out of the M-2remaining stages, choose the one that yields the smallest sum-of-squaredcoefficient values for the resulting partial filter #3, and repeat thisprocess for each of the stages until the sequence is complete, accordingto an embodiment.

In an embodiment, the distribution of stages occurring at block 3204also includes identifying those stages with the highest sum of squaredcoefficient values and distributing those stages among the other stagesin the cascade. In one example, those stages with the highest sum ofsquared coefficient values may be spaced apart equally from one anotherin the cascade.

VIII. EXEMPLARY COMPUTER SYSTEM

Embodiments of the invention may be implemented using hardware,programmable hardware (e.g., FGPA), software or a combination thereofand may be implemented in a computer system or other processing system.In fact, in one embodiment, the invention is directed toward a softwareand/or hardware embodiment in a computer system. An example computersystem 3300 is shown in FIG. 33.

Computer system 3300 includes one or more processors (also calledcentral processing units, or CPUs), such as a processor 3304. Processor3304 is connected to a communication infrastructure or bus 3306. In oneembodiment, processor 3304 represents a field programmable gate array(FPGA). In another example, processor 3304 is a digital signal processor(DSP).

One or more processors 3304 may each be a graphics processing unit(GPU). In an embodiment, a GPU is a processor that is a specializedelectronic circuit designed to rapidly process mathematically intensiveapplications on electronic devices. The GPU may have a highly parallelstructure that is efficient for parallel processing of large blocks ofdata, such as mathematically intensive data common to computer graphicsapplications, images and videos.

Computer system 3300 also includes user input/output device(s) 3303,such as monitors, keyboards, pointing devices, etc., which communicatewith communication infrastructure 3306 through user input/outputinterface(s) 3302.

Computer system 3300 also includes a main or primary memory 3308, suchas random access memory (RAM). Main memory 3308 may include one or morelevels of cache. Main memory 3308 has stored therein control logic(i.e., computer software) and/or data.

Computer system 3300 may also include one or more secondary storagedevices or memory 3310. Secondary memory 3310 may include, for example,a hard disk drive 3312 and/or a removable storage device or drive 3314.Removable storage drive 3314 may be a floppy disk drive, a magnetic tapedrive, a compact disc drive, an optical storage device, tape backupdevice, and/or any other storage device/drive.

Removable storage drive 3314 may interact with a removable storage unit3318. Removable storage unit 3318 includes a computer usable or readablestorage device having stored thereon computer software (control logic)and/or data. Removable storage unit 3318 may be a floppy disk, magnetictape, compact disc, Digital Versatile Disc (DVD), optical storage disk,and/any other computer data storage device. Removable storage drive 3314reads from and/or writes to removable storage unit 3318 in a well-knownmanner.

Secondary memory 3310 may include other means, instrumentalities, orapproaches for allowing computer programs and/or other instructionsand/or data to be accessed by computer system 3300. Such means,instrumentalities or other approaches may include, for example, aremovable storage unit 3322 and an interface 3320. Examples of theremovable storage unit 3322 and the interface 3320 may include a programcartridge and cartridge interface (such as that found in video gamedevices), a removable memory chip (such as an EPROM or PROM) andassociated socket, a memory stick and universal serial bus (USB) port, amemory card and associated memory card slot, and/or any other removablestorage unit and associated interface.

Computer system 3300 may further include a communication or networkinterface 3324. Communication interface 3324 enables computer system3300 to communicate and interact with any combination of remote devices,remote networks, remote entities, etc. (individually and collectivelyreferenced by reference number 3328). For example, communicationinterface 3324 may allow computer system 3300 to communicate with remotedevices 3328 over communications path 3326, which may be wired and/orwireless, and which may include any combination of local area networks(LANs), wide area networks (WANs), the Internet, etc. Control logicand/or data may be transmitted to and from computer system 3300 viacommunication path 3326.

In an embodiment, a tangible apparatus or article of manufacturecomprising a tangible computer useable or readable medium having controllogic (software) stored thereon is also referred to herein as a computerprogram product or program storage device. This includes, but is notlimited to, computer system 3300, main memory 3308, secondary memory3310, and removable storage units 3318 and 3322, as well as tangiblearticles of manufacture embodying any combination of the foregoing. Suchcontrol logic, when executed by one or more data processing devices(such as computer system 3300), causes such data processing devices tooperate as described herein.

In another embodiment, the invention is implemented primarily inhardware using, for example, hardware components such as applicationspecific integrated circuits (ASICs), stand alone processors, and/ordigital signal processors (DSPs). Implementation of the hardware statemachine so as to perform the functions described herein will be apparentto persons skilled in the relevant art(s). In embodiments, the inventioncan exist as software operating on these hardware platforms.

In yet another embodiment, the invention is implemented using acombination of both hardware and software. Field-programmable gatearrays (FPGA) could, for example, support such an embodiment.

IX. CONCLUSION

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. It will be apparent to persons skilled inthe relevant art(s) that various changes in form and detail can be madetherein without departing from the spirit and scope of the invention.Thus the present invention should not be limited by any of theabove-described exemplary embodiments, but should be defined only inaccordance with the following claims and their equivalents.

What is claimed is:
 1. A filter, implemented in hardware, configured toreceive an input signal and generate an output signal, the filtercomprising: a plurality of first stages, each stage of the plurality offirst stages having an order of four or greater; and one or more secondstages, each stage of the one or more second stages having an order lessthan four, wherein the one or more second stages and the plurality offirst stages are coupled together in cascade and wherein a total orderof the plurality of first stages is higher than a total order of the oneor more second stages.
 2. The filter of claim 1, wherein each stage ofthe plurality of first stages includes at least two multipliers.
 3. Thefilter of claim 1, further comprising a multiplier positioned at the endof a last stage in the cascade.
 4. The filter of claim 1, furthercomprising a scaling multiplier disposed between a pair of stages in thecascade.
 5. The filter of claim 4, wherein the scaling multiplier isconfigured to multiply a signal received at the scaling multiplier by apower of two.
 6. The filter of claim 5, wherein the scaling multiplieris further configured to shift a number of bits associated with anoutput of a previous stage to the right if the power of two is less than1, and to shift the number of bits associated with the output of theprevious stage to the left if the power of two is greater than
 1. 7. Thefilter of claim 4, wherein a value of the scaling multiplier is chosento bring a DC gain of a transfer function representing all previousstages closer to
 1. 8. The filter of claim 1, further comprising aregister or delay block located after one or more stages within thecascade.
 9. The filter of claim 1, wherein the filter is a digitalfilter.
 10. The filter of claim 9, wherein the digital filter is afinite impulse response (FIR) filter.
 11. The filter of claim 9, whereinthe digital filter is an infinite impulse response (IIR) filter.
 12. Thefilter of claim 1, wherein the filter is an analog filter.
 13. Thefilter of claim 1, wherein each stage of the plurality of first stagesis not identical to each other.
 14. A method, performed by a processingdevice, of determining factors of a filter, the method comprising:organizing pairings of angle values into pairing candidates; defining athreshold to indicate an upper bound on the number of pairingcandidates; exchanging a first pairing candidate above the thresholdwith a second pairing candidate below the threshold; generating a matrixbased on the pairing candidates below the threshold; determining alowest predicted total quantization cost between all pairing candidatesrepresented within the matrix; and using the pairing candidates thatresult in the lowest predicted total quantization cost to determine thecoefficients of the filter.
 15. The method of claim 14, wherein a givenpairing of angle values is represented by θp and θq, and a firstcoefficient of a given stage of the filter is provided as −2·(cos θp+cosθq) while a second coefficient of the given stage of the filter isprovided as 2·(1+2·cos θp·cos θq).
 16. The method of claim 14, whereinthe organizing comprises rearranging the pairings of angle values from alowest quantization cost to a highest quantization cost.
 17. The methodof claim 16, wherein the quantization cost for a given pairing of anglevalues is associated with a number of bits required to represent thegiven pairing of angle values within a stage of the filter.
 18. Themethod of claim 17, wherein the number of bits does not include a signbit.
 19. The method of claim 14, wherein the exchanging comprisesexchanging a first pairing candidate greater than the threshold with asecond pairing candidate below the threshold, wherein a quantizationcost of the first pairing candidate is the same as a quantization costof the second pairing candidate.
 20. The method of claim 14, wherein thedetermining comprises using Binary Integer Programming to determine abinary vector x that minimizes a linear cost function.
 21. The method ofclaim 20, wherein the binary vector x is constrained by the inequalityAx≦b, where A represents the matrix and b represents a vector consistingof ‘1’ values.
 22. The method of claim 20, wherein the linear costfunction is represented by f^(T)x, where f^(T) is a cost vector.
 23. Themethod of claim 20, wherein one or more entries of the cost vector arerelated to one or more quantization costs among pairing candidates. 24.The method of claim 20, further comprising updating the entries of thecost vector in a feedback loop until the actual total quantization costis substantially the same as the predicted total quantization cost. 25.The method of claim 14, wherein the pairings of angle values comprisenatural pairs of filter zeros.
 26. The method of claim 25, wherein thenatural pairs of filter zeroes comprise 2^(nd) order factors.
 27. Themethod of claim 26, wherein the natural pairs of filter zeroes comprise4^(th) order factors.
 28. The method of claim 14, further comprising:after determining the coefficients of the filter for a given threshold,defining a different threshold to indicate a different upper bound onthe number of pairing candidates and repeat the exchanging, generating,determining and using with the different threshold.
 29. A method,performed by a processing device, for determining a sequence of aplurality of stages of a filter, the method comprising: determining asum of squared coefficient values for each stage of the plurality ofstages of the filter; arranging the plurality of stages of the filter incascade such that: a first stage position in the cascade includes astage having a lowest sum of squared coefficient values among each stageof the plurality of stages, and a subsequent stage position includesanother stage of the plurality of stages, such that a partial filtercomprising the another stage and each previous stage in the cascade hasa lowest sum of squared coefficient values among the possible stages tochoose for the another stage.
 30. The method of claim 29, furthercomprising: identifying one or more stages in the arranged cascadehaving the highest sum of squared coefficient values among all of thestages in the arranged cascade; and distributing the one or more stageshaving the highest sum of squared coefficient values among the otherstages in the arranged cascade.
 31. The method of claim 30, wherein thedistributing comprises distributing such that the one more stages havingthe highest sum of squared coefficients are substantially spaced equallyapart from one another in the cascade.
 32. The method of claim 29,wherein more than half of the stages of the plurality of stages eachhave an order of four or greater.
 33. The method of claim 29, wherein asubset of the plurality of stages each have an order of four or greater,and the total order of the subset of the plurality of stages is greaterthan a total order of a remainder of the stages in the plurality ofstages.